Frequency modulated ballast with loosely coupled transformer for parallel gas discharge lamp control

ABSTRACT

A frequency dependent controller utilizes a loosely coupled transformer, adapted to change the operating characteristics of the transformer by varying its operating frequency in accordance with the requirements of a load to be supplied with power. The controller is particularly adapted for dimming and adjustably controlling a set of gas discharge lamps such as fluorescent lamps, connected in parallel such that failure of one lamp does not affect operation of the other lamps in the set.

CROSS REFERENCE TO RELATED APPLICATIONS

This is a continuation-in-part of copending U.S. patent application Ser.No. 09/063,934, filed Apr. 21, 1998 now U.S. Pat. No. 6,088,249,entitled “Frequency Modulated Ballast With loosely Coupled Transformer”which is a continuation-in-part of copending U.S. patent applicationSer. No. 08/982,974 filed Dec. 2, 1997 now U.S. Pat. No. 5,933,340entitled, “Frequency Controller for a Loosely Coupled Transformer Havinga Shunt with a Gap and Method Therefor,” which is hereby incorporated byreference in its entirety.

BACKGROUND OF THE INVENTION

1. Field of the Invention

The present invention relates to a dimmable electronic ballast for gasdischarge lamps and other devices, and more particularly to a dimmableelectronic ballast for lamps comprising a loosely coupled transformer,in which current to a plurality of parallel connected gas dischargelamps is adjusted to dim the illumination provided by the lamps.

2. Description of the Related Art

An electronic ballast is used to convert commercial electrical powerinto a high voltage electrical signal sufficient to create and maintaina plasma in a gas discharge lamp. Typical examples of gas dischargelamps are fluorescent lamps and neon lamps. Transformers are typicallyused in ballast circuits to boost the voltage across the lamps.

Shunted transformers (also generally known as loosely coupledtransformers, or leakage transformers) are transformers having higherthan usual leakage inductance. Where the intent in standard powertransformers is to produce a transformer that is very tightly coupled,that is to create a transformer with very low reactance, the opposite istrue with respect to shunted transformers where the reactance resultingfrom a shunt is traded off for current limiting capability. The geometryof the transformer core, and the nature of the windings of the coils,are factors that can affect the leakage inductance. All knowntransformer designs have at least a small amount of reactance, thus theconcept of a leakage transformer is based on a relative scale.

As the terms are used herein, a “tightly coupled” transformer isconsidered to be one in which a very high percentage of the magneticflux developed in the transformer s primary winding is delivered to itssecondary winding. See pages 223, 224, 234, 235 and other generalinformation in Electronic Transformers and Circuits, 2nd Edition, byReuben Lee, published in 1955 by John Wiley, New York, N.Y. For example,placing the primary winding of a transformer on top of its secondarywinding or interleaving the windings will provide a tightly coupledtransformer in which substantially all the flux developed in the primarywinding “flows” in the secondary winding by physical definition.

A “loosely coupled” transformer, on the other hand, is considered to bea transformer in which a lesser amount of the magnetic flux developed inthe transformer's primary winding finds its way to the secondarywinding. This relationship can also be expressed in terms of atransformer's coupling ratio, as defined by Lee on page 235 of hisaforementioned reference, where “k”, the coefficient of coupling (whichvaries from 0 to 1) is determined from:

(I₂ ²Z₂/E₁I₁)_(MAX) =k ²/2(1+(1−k ²)^(½) −k ²  (1)

in which, “I” indicates current, “E” indicates voltage, “Z” indicatesimpedance, the “1” subscript refers to the primary and the “2” subscriptrefers to the secondary of a transformer under consideration. Ratiovalues for “k” below 0.90 are considered to be loosely coupled withratio values above 0.99 (an arbitrary dividing line) considered to betightly coupled.

In addition to using magnetic flow as a measure of coupling, it is alsopossible to ascertain coupling using the inductance exhibited by theprimary when the secondary is open and when it is shorted. The conditionof the secondary winding circuit, open or shorted, determines the amountof current flow and, derivatively, the inductance exhibited by theprimary winding. The ratio of the primal winding inductance under thesetwo extremes gives rise to another form of coupling measurement as shallhereinafter be further described.

Leakage transformers may be gapped or ungapped, depending on the overalldesign and electrical characteristics. Loosely coupled shuntedtransformers having an air gap in one of their legs function in thefollowing fashion. Typically, a E-shaped or multi-legged magnetic coreis employed with the air gapped leg having a specific magneticreluctance determined in part by the size of the air gap. One of thenon-shunted legs holds the primary winding and the other non-shunted legholds one or more secondary windings. The shunt is not usually providedwith a winding.

The loop including the gapped leg has a fixed reluctance that issignificantly higher than that of the secondary loop when the secondaryis at low load or is entirely unloaded. In fact, at low load, thesecondary winding magnetic loop will have most of the flux flowingthrough it and the secondary voltage will be high. As the loadincreases, the reluctance of the secondary loop increases and thesecondary voltage decreases. As the secondary load approaches a short oris actually shorted (secondary voltage is zero), the majority ofmagnetic flux now flows through the gapped leg as its magneticreluctance is lower than the high reluctance of the secondary windingloop. Thus, at low secondary voltage, the current is high, but limitedto a value determined by the reluctance of the gapped leg.

There are basically two types of variable leakage reactance transformersused to control or limit current flow. As described in U.S. Pat. No.4,123,736 to Brougham entitled LEAKAGE REACTANCE TRANSFORMER, they arecommonly referred to as the moving coil type and the moving shunt type.The moving coil type transformer relies on moving one of its windingsrelative to the other to adjust leakage reactance. In the shunted typetransformer, a steel shunt is movably mounted on a frame located betweenspaced apart primary and secondary windings and is moved into and out ofthe space between the windings to vary the transformer's reactance. Inboth types of transformers, degrees of control are predicated onmechanical movement of a winding or a shunt and it is, therefore,difficult to achieve precision current control at fixed frequency.Further, as noted in the Brougham reference, the costs of suchtransformers is relatively high, especially when higher costarrangements are needed to overcome problems presented by wear, jammingand the lack of precision control.

In U.S. Pat. No. 4,187,450 to Chen for HIGH FREQUENCY BALLASTTRANSFORMER, a transformer is described that is particularly useful inconjunction with solid state, high frequency push-pull inverters forsupplying power to discharge lamps. The Chen transformer comprises apair of facing E-shaped core sections disposed adjacent to one anotherin a mirror image fashion with their corresponding legs aligned but withan air gap provided between the middle, non-touching legs of the core.The transformer is described as being wound in a special fashion toovercome prior art limitations of insufficient ballasting reactance(needed to overcome the negative impedance at startup exhibited by gasdischarge lamps) and magnetic leakage. This reference, however, does notteach any method of utilizing frequency or current control to regulatepower or signals provided to the secondary of a loosely coupledtransformer.

Another air gapped transformer is described in U.S. Pat. No. 4,888,527to Lindberg for REACTANCE TRANSFORMER CONTROL FOR DISCHARGE DEVICES. Inthis prior art device for obtaining current limited control of gasdischarge lamps, one leg of a three legged transformer is provided withan air gap and fixed reluctance. The transformer's reactance is variedby means of a separate control winding that varies the reluctance of thetransformer leg on which it is wound as a function of a variableimpedance included in a control circuit used to drive the controlwinding.

U.S. Pat. No. 5,192,896 to Qin for VARIABLE CHOPPED INPUT DIMMABLEELECTRONIC BALLAST teaches an output transformer having a looselycoupled primary and secondary winding and a pair of slidable magneticshunts. The Qin transformer is constructed from a pair of facingE-shaped ferrite cores having an air gap in its center leg. The primaryand secondary windings are separated from each other by a pair of shunthousings in which the movable shunts are slidably mounted. By adjustingthe position of the shunts, the parameters of the transformer can beadjusted to match the load requirements.

As described above, there were a number of prior art transformerarrangements that sought to take advantage of the inherentcharacteristics of shunted transformers by varying winding methods orpositioning, using slidable shunts and adding control windings tovarious portions of such transformers. While these attempts at improvingthe results achieved by control or modification of reactancetransformers did achieve better operating results or manufacturingcosts, they still failed to yield the degree of precision, low cost,efficiency and versatility required by modern power transferringarrangements.

Co-pending U.S. patent application entitled “Frequency Controlled, Quickand Soft Start Gas Discharge Lamp Ballast and Method Therefor,” Ser. No.08/982,975, describes an electronic ballast, and is hereby incorporatedby reference in its entirety.

The brightness of a gas discharge lamp can be controlled by adjustingthe output power of the ballast. Dimmable electronic ballasts typicallyuse pulse width modulation (PWM) to control output power. In a typicalPWM circuit, the width of a square wave pulse is adjusted so as tochange the total power delivered to the load. It would be undesirable,in many designs, to vary the frequency because many ballast designs havea resonant output stage that helps boost the output voltage. The drivingfrequency for the output stage, including the transformer, of a PWMcircuit is typically held constant to maintain the resonance.

It would therefore be desirable to have an electronic ballast designthat is efficient and less expensive than current designs. It would alsobe desirable to have an electronic ballast design that does not requirea resonant circuit in its output stage.

SUMMARY OF THE INVENTION

The present invention improves on the prior art devices by providing asimple and inexpensive dimmable electronic ballast that takes advantageof the properties of a loosely coupled transformer. The circuit of thepresent invention employs frequency modulation in combination with theelectrical properties of the loosely coupled transformer to control theelectrical current through the load and uniquely connects a plurality ofgas discharge lamps in true parallel such that failure of one lamp doesnot affect operation of the remaining gas discharge lamps. In addition,by varying the frequency in the primary coil of the loosely coupledtransformer powering the lamps, the current can be appropriately limitedin the secondary coil through the inherent reactance of the transformerand the light output of the lamps controllably dimmed.

BRIEF DESCRIPTION OF THE DRAWINGS

The present invention is explained in the context of use within anelectronic ballast. A more complete understanding of the presentinvention can be obtained by considering the following detaileddescription of the preferred embodiments thereof in conjunction with theaccompanying drawings, in which:

FIG. 1 is a block diagram of the main functional elements of a dimmingelectronic ballast in which transformer control is employed.

FIG. 2 is a schematic diagram of a dimming electronic ballast wherein aloosely coupled transformer under frequency control is used to supplypower to a load comprising gas discharge lamps.

FIG. 3 illustrates the output voltage haversines obtained from a fullwave rectifier bridge utilized in the electronic ballast shown in FIG.2.

FIG. 4 depicts a schematic diagram of a non-dimming electronic ballastwherein a loosely coupled transformer under frequency control is used tosupply power to a load comprising gas discharge lamps.

FIG. 5(a) is a diagram of a loosely coupled transformer of the typeincorporated in embodiments of the present invention.

FIG. 5(b) is a diagram of an alternative gapless embodiment of a looselycoupled transformer.

FIG. 6 shows an equivalent model circuit of a loosely coupledtransformer.

FIG. 7 depicts a plot of winding current in a loosely coupledtransformer as a function of the frequency of the voltage appliedthereto.

FIG. 8 shows a block diagram of a control arrangement which utilizes aloosely coupled transformer.

FIG. 9 schematically depicts use of a loosely coupled transformer underfrequency control to supply power to a thermoelectric load in accordancewith the present invention.

FIG. 10 shows a block diagram of the use of a loosely coupledtransformer under frequency control to supply a train of pulses to aremote receiving circuit.

FIG. 11 schematically illustrates use of a loosely coupled transformerunder frequency control in accordance with the present invention tosupply power to a sub-station or industrial.

FIG. 12 is an electrical schematic diagram of another embodiment of adimmable electronic ballast.

FIG. 13 is a block diagram schematic of a feedback control loop.

FIG. 14 is an electrical schematic diagram of an embodiment of adimmable electronic ballast according to the present invention forcontrol of parallel connected gas discharge lamp fixtures.

FIG. 15 is an electrical schematic diagram of the gas discharge lampsconnected to the ballast shown in FIG. 14.

FIG. 16 is a simplified electrical schematic diagram of the output driveportion of the embodiment shown in FIG. 14 for remote installation ofthe loosely coupled transformers T2, T3, T4, and T5 in accordance anembodiment of the present invention.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS

In the preferred embodiments of the invention, the dimming ballast forparallel gas discharge lamps is a combination of the dimming electronicballast, whose preferred embodiments are described in the parent U.S.patent application Ser. No. 09/063,934, filed Apr. 21, 1998, entitled“Frequency Modulated Ballast With loosely Coupled Transformer,” withparallel, loosely coupled transformer circuits illustrated in FIGS. 14,15 and 16. A reader familiar with the dimming electronic ballastdescribed in the parent application should skip to the heading, “DimmingBallast For Parallel Gas Discharge Lamps” for a detailed description ofthe combination making up the preferred embodiments of the presentinvention.

Frequency Modulated Ballast With Loosely Coupled Transformer:

Referring now to the drawings, wherein like reference numerals have beenused in the several views to identity like elements. FIG. 1 illustratesa block diagram of an electronic dimming ballast 2. In its most generalform, the electronic dimming ballast 2 may be used to control variousloads, and as such the ballast 2 may also be referred to herein as acontroller. The general nature of the controller is a FrequencyModulated (FM) circuit that utilizes the inherent current limitingcharacteristics of transformer 178 to control power to a load.Variations in frequency will control the current flow through thesecondary winding of the transformer 178, and thereby limit theelectrical power to a load.

One of the reasons that FM is applicable in this circuit is that thereis no attempt to maintain a resonance in the output stage. Many priorcontrollers attempt to maintain a resonant stage, which typicallyrequires operation within a narrow frequency band. While the presentcontroller may have circuit dynamics with a resonant frequency, there isno attempt to excite and maintain such a resonance.

The load is shown as a lamp 4 in FIG. 1, but the load may also be anelectric motor, heater, cooler, or another device as would be known toelectrical engineers. The design is also applicable to current limitingDC to DC converters and AC to DC converters.

The controller comprises a power supply circuit, which may include arectifier, amplifier, power factor corrector, filter, buffer, and othersignal processing circuitry that is generally used in power supplycircuits adapted for use with commercial Alternating Current (AC) power.The controller also comprises a Voltage Controlled Oscillator (VCO),shown as 11 in FIG. 1, connected to the power supply circuit, to producea frequency modulated (FM) output signal in response to a voltage input.The controller further comprises a loosely coupled transformer 178having at least one primary winding and at least one secondary winding.The primary winding is electrically connected to the adjustablefrequency output signal, in addition to being connected to otherelements of the power supply. The secondary winding is electricallyconnected to supply current to the load. The transformer 178 has thegeneral property shown in FIG. 7, wherein the current in the secondarywinding is responsive to the adjustable frequency output.

A further description of FIG. 1 follows. A line voltage source 6, havinga frequency and voltage level normal for the specific location of theworld at which the ballast is being used, provides power to the ballast2. The supplied voltage is rectified by circuit 5 to a continuous seriesof haversines, see FIG. 3, and passed on to a voltage booster and powerfactor correction circuit 7. The circuit elements of block 7 serve toraise the voltage level to a self-sustaining value appropriate forcontrol circuit 9 and the gas discharge type of lamps 4 being used.Control circuit 9 forwards a reference voltage signal to the frequencycontrol circuit 11, which adjusts the frequency of its output inaccordance with that reference, subject to any changes in that outputfrequency called for and required by the dimming control circuit 15through its desired dimming level signal 17. As shown in FIG. 1, thedimming control circuit 15 also receives a brightness feedback signal 19from the lamps 4 which it compares to the desired dimming level signal17 to derive a differential signal, whenever signals 17 and 19 aredifferent, that is used to adjust the frequency of the voltage output byfrequency control circuit 11.

The output from frequency control circuit block 11 is fed to controlcircuit 9 where it is used to produce a closely matched square wavevoltage that is forwarded to transformer drivers circuit 13. Thetransformer drivers thereby created in block 13 are used to generatedrivers for a special transformer 178, to be described hereinafter ingreater detail, that is coupled to the lamps 4 whereby power isdelivered to the lamps 4.

FIG. 2 illustrates in greater detail the components of a dimmingelectronic ballast 2 used for coupling a set of fluorescent lamps 4 tothe source of electrical power 6 at input terminals 8 a and 8 b. Thepower source in this instance is 110 volts AC at a frequency of 60hertz, the standard power conditions found in the United States. Thedepicted ballast 2, however, can accept input voltages in the range of90 to 300 volts AC at frequencies of 50 to 60 hertz or 140 to 450 voltsDC. This permits the electronic ballast to function satisfactorily, infull accordance with its specifications, in any country in the worldunder most power conditions that will be encountered.

Resister 10 is connected in series with input terminal 8 a and serves asa fuse or current limiting device. Resistor 10 represents a simple wayto protect against overloads or excessive transients that wouldotherwise harm the ballast or compromise safety. Inductors 12 andcapacitors 14 and 16 form an Electromagnetic Interference (EMI) orcommon mode choke filter that reduces EMI conducted to terminals 8 a and8 b by limiting high frequency signals and passing only signals thathave a complete path through the ballast 2.

Full wave rectifier bridge 18 converts the input signal in standardfashion into the rectified output voltage signal 20 as shown in FIG. 3.The output voltage 20 of bridge 18 passes through several paths, thefirst including boost choke 22 a, diode 24 which provides power tooutput FETs 158 and 160, as will be discussed hereinafter, capacitor 26,resistor 30, diode 28 and capacitors 32 and 34. Capacitors 32 and 34 areconnected to input pin 36 of power factor chip 38.

Power factor chip 38 is a Motorola MC33262 integrated circuit chip whichis more fully described at pages 3-455 to 3-457 of Motorola'sAnalog/Interface ICs, Device Data, Volume 1, Revision 5, 1995.Integrated circuit chip 38 is a high performance, current mode powerfactor controller that is designed to enhance poor power factor loads bykeeping the AC line current sinusoidal and in phase with the linevoltage. Proper power factor control keeps the apparent input powerphase to ballast 2 close to that of the real power it consumes therebyincreasing the ballast's operating efficiency in that respect.

When power is turned on, the first half-cycle or haversine 40 thereofoutput by bridge 18, see FIG. 3, reaches a value of 170 volts peak withan input voltage of 120 volts rms. Capacitors 26, 32 and 34 areinitially discharged. Capacitors 26 and 32, when discharged appearshorted and when subjected to the initial haversine in an AC circuit,act as a voltage divider as determined by their respective capacitivevalues, for the initial haversine 40. In this instance, if the value ofcapacitor 26 is selected to be 24 microfarads and the value of capacitor32 is chosen to be 48 microfarads, then capacitor 26, in the absence ofother limits, would have two thirds of the peak of voltage 20 or 113.3volts impressed across it and capacitor 32 would have one third ofvoltage 20 or 56.7 volts impressed across it. If the input voltage is aslow as 90 volts, the voltage across capacitor 32 is still more than 30volts, which is sufficient to turn on power factor chip 38. At the sametime, resistor 30 limits the in-rush current to capacitor 26. As aresult, the voltage division preciously referred to across capacitors 26and 32 actually occurs between capacitor 26, resistor 30 and capacitor32, with resistor 30 serving the dual function of also limiting in-rushcurrent on the first-half cycle. Capacitor 34 would have little effecton the voltage divider aspects of capacitors 26 and 32 since it is verysmall in value compared to capacitors 26 and 32.

Ordinarily, one third of the peak of voltage 20, a voltage of 56.7 voltsin this instance, across capacitor 32 would be more than sufficient toburn it out. However, the combination of resistors 42 and 44 with FET 46imposes a pre-set voltage limit across capacitor 32, which also protectsIC chip 38. The values of resistors 42 and 44 are selected to have thevoltage limit across capacitor 32 set to 15 volts in this instance. Whenthe voltage across capacitor 32 exceeds its preset limit, FET 46 turnson pulling its base and line 48 to ground. Since line 48 is connected toone side of capacitor 26 and the base of diode 28, when line 48 goes toground, capacitor 26 acts as an open circuit to further haversines andbecomes a filter capacitor to the high DC voltage to power FETs 158 and160. At the same time, diode 28 is back biased thereby eliminating thefurther flow of current through diode 28 and resistor 30 and preventingcapacitor 32 from being charged above its preset limit of 15 volts. Thisis true regardless of the value of input voltage in its nominal range of90 to 300 volts AC, 50 to 60 Hertz.

In this manner, the circuit arrangement described above limits the totalin-rush current at an input of 120 volts, where the voltage and currentare 90 degrees out of phase with each other, the worst case, to 4.3 ampsat an input of 120 volts AC. At the same time, the circuit insures thatthe appropriate voltage is applied to pin 36 of the power factor chip 38to start it without ongoing loss of power even when the input voltage isat lowest expected value, in this case 90 volts AC. Further, even atthis lowest possible input voltage, the circuit elements cooperate toprovide an “instant on” capability for the power factor chip which getsturned on in the first half cycle of power over an input voltage rangeof 90 to 300 volts AC.

By way of comparison, the prior art method of using a resistor connectedbetween the output of a ballast's power supply and the voltage supplypin of power correction circuit, such as IC 38, would require a designtradeoff between power loss and turn on time over the expected inputvoltage range of 90 to 300 volt AC. For example, a 100k ohm resistorconnected to rectifier bridge 18 and across capacitor 32 at 300 volts ACinput would cause a current of about 4 milliamps, draw about 0.9 wattsand take approximately 0.25 seconds to charge capacitor 32 to theminimum level required to start IC 38. On the other hand, the same 100kohm resistor at 90 volts AC input would draw only 80 milliwatts, buttake almost 2.0 seconds to turn on IC 38. As will be explainedhereinafter, the availability of voltage to turn on the lamps isdependent upon operation of the power circuit chip 38. The longer IC 38takes to turn on, the longer it takes for the lamps to be illuminated.

Further, if the power factor chip is initially turned on by connecting aresistor from the output of a full wave rectifier bridge or other DCsource to the chip's input pin and a capacitor, such as capacitor 32,that resistor continues to dissipate wattage even when it is not neededafter IC 38 is started. This usually wastes, depending on the size ofthe resistor selected, from approximately 80 milliwatts to 900milliwatts and while that seems low when compared to other electronicdevices, the overall saving in a setting where many lamps and ballastsare used, such as in a parking garage or large warehouse with hundredsof lamp fixtures, provides a considerable cost saving.

It is important to note that once appropriate power to chip 38 isprovided, FET 46 is turned on as explained above, and as a result, diode28 and resistor 30 are removed from the active circuit path. Sincecurrent is now supplied on a boost basis through coil 22 b, diode 70 andresistor 72 to capacitor 32, resistor 30 isn't needed and the lack ofcurrent flow therethrough avoids power waste. The net effect is thatresistor 30 is used during the first half cycle of operation to limitin-rush current and act as part of a voltage divider until FET 46conducts after which it is effectively removed from the operationalportion of the ballast's power supply circuit.

In addition, since the time constant of the resistor capacitorcombination at input pin 36 limits voltage buildup across the capacitorand to the input pin, it takes longer for the voltage across thatcapacitor to reach an appropriate level to turn on the power factor chipand the ballast. Conversely, the above-described circuit elements of theballast insure that the ballast is turned on in the first half cycle ofoperation, especially for lower than usual input voltages, while that isnot true in the prior art.

Once initiated, IC chip 38 is made self-supporting through the combinedeffects of associated circuit elements which use the output voltage 20from rectifier bridge 18 to power chip 38 and the ballast controlcircuitry along with the lamps 4 with IC chip 38 serving to maintain theload power in phase with the input power. This is accomplished byforcing an inductive kick to occur in choke 22 a, and derivatively inits secondary coil 22 b, in phase with the haversines available at theoutput of full wave rectifier bridge 18. It should be noted that the ICchip 38 will shut down with no load because there will be no sustainingDC voltage.

The output voltage 20 from bridge 18 is passed to resistors 50 and 52,which form a voltage divider, and partially to capacitor 54 which helpsfilter voltage applied to input pin 56 of IC chip 38. Input pin 58 of ICchip 38 is held at ground. By appropriate choice of resistor values forresistors 50 and 52, the voltage on pin 56, which is in phase with thehaversines of bridge output voltage 20, is typically set at around 2volts peak. Internal to chip 38, that voltage is passed through a drivecircuit (not shown) to appear on output pin 60. That voltage is appliedto the gate of FET 62 and turns FET 62 on whenever the appropriatevoltage level is reached during each haversine.

When FET 62 turns on, it very quickly pulls the right side of coil 22 ato ground, and when it releases, causes an inductive kick in coil 22 aand a reflective inductive kick in coil 22 b, both of which are in phasewith the haversines derived from the input power line. The inducedvoltage level is determined by the voltage divider formed by resistors64 and 66. In this application, the values of resistors 64 and 66 areselected to produce a total value of 435 volts on capacitor 26. Thetotal voltage is the result of the bridge output voltage 20 and thevoltage resulting from the inductive kick in coil 22 for the time periodthat FET 62 is turned off. The 435 volts charge capacitors 26 and 68 tothat level in phase with the input line voltage, or nearly so with thevariance being approximately one decree out of phase. Capacitor 68 alsoserves as a high frequency filter.

Feedback from the junction of resistors 64 and 66 is provided to inputpin 76 of IC chip 38 as a reference voltage Vset which is used to informthe internal circuitry of IC chip 38 that the correct DC voltage hasbeen reached. When that occurs the drive voltage is removed from pin 60and FET 69 is permitted to turn off until the next haversine is present.

Since capacitor 26 may be drained when the ballast 2 is loaded, theresultant 435 volts is also used via feedback to keep IC chip 38 poweredon as well as to provide power to the remainder of the ballast 2 and tothe lamps 4. Feedback power to IC chip 38 is provided by coil 22 b,through diode 70 and resistor 72 to capacitors 32 and 34. Capacitor 32applies power to input pin 58 of IC 38 as previously described throughthe voltage limiting combination of resistors 42 and 44 and FET 46.Diode 70 and resistor 72 are inserted in the path from coil 22 b tocapacitors 32 and 34. Diode 70 serves to prevent discharge fromcapacitors 32 or 34 from causing unwanted current flow into coil 22 bwhile resistor 72 serves to limit current in that circuit leg tocapacitors 32 and 34 and acts as a filter in conjunction with capacitors32 and 34 for pin 58.

The combination of resistors 78 and 80 and capacitor 82 serve to protectFET 62 by limiting and filtering the current signal drawn when FET 62turns on. Resistor 80 is selected to have a low value, typically a fewtenths of an ohm. Drive voltage is supplied to output pin 88 andresistor 80 serves to limit that to a current value that can easily betolerated by FET 62. In addition, the voltage drop across resistor 80determines the level at which FET 62 is turned off since it provides adelay that results from its RC combinational effect with capacitor 82.In addition, resistor 78 and capacitor 82 provide a high frequencyfilter capability for current flowing to pin 88.

On balance, the net effect of the front end portion of ballast 2 is toprovide a precision, high voltage DC power supply with power factorcompensation for an electronic ballast that turns on in its first halfcycle of operation at reduced power consumption even when the inputvoltage level is at its lowest expected value. The front end portionalso features a limited in-rush current capability that is automaticallyremoved from the operational portion of the circuit thereby conservingpower. The front end ballast portion also provides isolated controlcircuit power. In fact, the use of transformers to power the lamps 4insures that there is no direct electrical connection from the inputpower source to the control circuit or the output (the lamps).

Current flowing through coil 22 a causes current to flow in secondarycoil 22 c by transformer action. When coil 22 a experiences thepreviously described inductive kick, a like and proportional increase incurrent flow is experienced in coil 22 c. By selecting an appropriateturns ratio between coils 22 a and 22 c, the induced voltage in coil 22c can be set to any desired level. In this case, that level can be aslow as 14 volts and as high as 40 rectified volts DC, depending on thecontribution of the inductive kick to voltage generated in coil 22 c andthe load being driven by the ballast 2. As indicated by the dots in FIG.2 alongside coil 22 a, 22 b and 22 c, their phases are chosenaccordingly.

The power for the control circuit portion 2 of ballast 2 is derived fromcoil 22 c and passes through diode 90, which prevents reverse currentflow into the front end portion 2 of ballast 2, to input pin 92 ofintegrated circuit chip 94, a Texas Instruments TPS2813 multi-functionchip. The voltage is filtered for high frequency by capacitors 96 and98. Input pin 92 serves as the input to an internal voltage regulator inIC chip 94. The output of that regulator is pin 100 which is held at aconstant value of 11.5 volts by the regulator's action over the range ofthe 14 to 40 volts which appears on input pin 92. Output pin 100 isconnected to input pin 102 of the CMOS integrated circuit chip 104, anRCA voltage controlled oscillator. Output pin 100, in the dimmingversion of ballast 2, is also connected to operational amplifier 106, anLMC6032.

Control of IC chip 104 is based on the capacitive value of capacitor 116which is connected across pins 112 and 114 of IC chip 104 and the valuesof resistors 122 and 124. The value of resistor 122 determines the lowerfrequency operating limit of IC chip 104 while the value of resistor 124determines its upper frequency limit. The voltage on input pin 126determines the operating frequency of the output voltage, a DC squarewave, on pin 150 of IC chip 104. If the voltage on pin 126 is zerovolts, the output on pin 150 oscillates at its lowest frequency asdetermined by resistor 122. If the voltage on pin 126 reaches it highestvalue, then the output on pin 150 of chip 104 oscillates at its highestpossible frequency as set by resistor 124.

As noted above, the values of capacitor 116 and resistors 122 and 124determine the minimum to maximum frequency range of response for chip104. If the voltage on pin 126 is zero volts, for example, then thevalues of resistor 122 and capacitor 116 determine the minimum frequencyof the voltage on pin 150. If the voltage on pin 126 reaches its maximumvalue of V_(cc), the voltage on pin 102, then the values of resistor 124and capacitor 116 determine the maximum frequency of the square wavepresent on pin 150. Further, it should be recognized that the frequencycontrol range is linear, that is, for example, a 10% change in voltageon pin 126 will produce a 10% change in the frequency of the voltage onpin 150. Alternatively stated, the values of resistors 122 and 124 alsodetermine the slope of the frequency range from its minimum to maximumvalues.

A set of matched resistors 128 is coupled between operational amplifiers106 and 110 and a dimming reference voltage source 130 that is comprisedof current sensing coil 132, resistor 134 and diode 136. Resistor set128 is a set of equivalent valued resistors that are matched to atolerance of 50 parts per million which creates a very good differentialamplifier when used in conjunction with amplifiers 106 and 110. As usedwith resistor set 128, the resultant differential amplifiers 106 and 110have a very high common mode rejection ratio which is important sincethe lines going out to the dimming control may run long distances andthe resulting voltage variations, if slight, will need to be accountedfor by the ballast control circuitry 2 b.

Voltage from a dimming switch (not shown) that is located near the lampsto be controlled, is applied across terminals 138 and 140. That voltageis applied through resistors in set 128 to the inputs of differentialamplifier 106. The output of amplifier 106 is fed again through one ofthe set 128 resistors to a summing point 141 and from there to thepositive input I of differential amplifier 110. At the same time, areference current induced by current in the high voltage lines to one ofthe lamps which is on or illuminated, is derived by transformer actionin coil 132 and forwarded to a resistor in set 128 and the resultantvoltage is also passed to summing point 141 and input 108 b of amplifier110. The actual voltage on input pin 108 b is, in this case, the averageof the dimmer voltage and the lamp reference voltage as derived fromcoil 132. If the dimmer voltage is assumed to be 3 volts and the lampreference voltage is assumed to be 3.5 volts, then the average voltageis 3.25 volts and that is what is applied to input 108 b of amplifier110.

The output of amplifier 110 is fed through resistor 142 to input pin 126of chip 104 thereby changing the voltage on pin 126 and the operatingfrequency of the voltage on output pin 150. This will cause a change inthe voltage applied to and the brightness of the lamps 4, raising orlowering the reference current developed in coil 132 as desired. Whenthe brightness called for by the voltage across terminals 138 and 140 isreached, the output of differential amplifier 110 no longer changes fromthe value called for by the dimmer and pin 126 is then left at aconstant voltage. In the above example, this would mean that the voltagefrom the dimmer is at 3 volts and the lamp reference voltage is also 3volts. That makes their sum 3 volts which holds pin 126 of chip 104constant, that is, until a change in brightness is called for.

A simplified block diagram of a negative feedback control loop, such asthe one embodied in FIG. 2, is shown in FIG. 13. Electrical current,depicted diagrammatically as flowing in line 1314, is measured by acurrent measuring device 1300. Examples of current measuring device 1300are a coil or transformer, and one example is shown as element 132 inFIG. 2. Device 1300 generates a signal, typically a voltage, depicteddiagrammatically as flowing in line 1302, that is connected to asummation circuit 1306 which is preferably an op-amp. A reference signalsource 1304 is preferably a potentiometer connected to a voltage source.Source 1304 produces a reference signal 1305, preferably a voltage, thatis connected to the summation circuit 1306. In a dimmable ballastembodiment, the reference voltage 1305 is typically set by a user toadjust the brightness of the lamps 4, which is shown as load 1320 inFIG. 13. In an alternative embodiment, the reference voltage 1305 may bea fixed value that is indicative of a desired brightness setting.

The summation circuit 1306 subtracts the signal 1302 from the referencesignal 1305 to arrive at a command signal 1308 that is connected to theinput of the voltage controlled oscillator 1310. The output 1312 of thevoltage controlled oscillator 1310 has its frequency adjusted inaccordance with the command voltage 1308. The frequency controlledoutput 1312 is used to drive a power supply circuit having a leakagetransformer, shown as block 1316 in FIG. 13. The leakage transformer,shown as 500 in FIG. 5, limits current based on the frequency of theoutput 1312, thus limiting current to the load 4. Thus, line 1318 inFIG. 13 represents a current limited electrical signal that is connectedto a load 1320, which is equivalent to the load 4 for the case of gasdischarge lamps. Thus, the voltage controlled oscillator 1310, theleakage transformer 500, the load 4, and the current measurement device1300 comprise parts of a feedback control system to maintain a desiredcurrent in the load 4.

In an alternative embodiment, shown in FIG. 12, the feedback control iseliminated in favor of an open-loop control system. The open-loopembodiment does not have the functional equivalent of block 1300 in FIG.13. Instead, control of the voltage controlled oscillator, 1310 in FIG.13, is based on the reference signal 1305 without benefit of thefeedback reference 1302. In many applications, an open-loop embodimentgives adequate performance and may be preferable due to its lower cost.

The embodiment shown in FIG. 12 has a similar electronic circuitarchitecture to the embodiment shown in FIG. 2. In FIG. 12, part CD4046is a voltage controlled oscillator, such as described in block 1310 ofFIG. 13. Components T1, Q3, and Q4 in FIG. 12 comprise a standardpush-pull driver circuit that is part of the power supply in block 1316of FIG. 13. The loosely coupled transformer in 1316 is shown ascomponent T2 in FIG. 12 and T2A in FIG. 2. As can be observed from FIG.12, there is no feedback from the secondary side of transformer T2.

A single ballast in this embodiment supplies from one to four lamps. Ina closed loop feedback embodiment, such as chosen in FIG. 2, theimportance of the negative feedback action of the lamp reference signal1302 is that the control system, shown in FIG. 13 places the appropriatevoltage on all lamps to create the brightness called for by the dimmerregardless of variations from ballast to ballast or dimmer switch todimmer switch. In essence this means that the same current in each lampis controlled by a single ballast in response to the feedback resultingfrom the sum of the dimmer voltage 1305 and the reference voltage 1302.This removes the effects of component variations from ballast to ballastand predicates lamp brightness under dimmer control on responsiveness tothe current feedback from coil 132, in the embodiment of FIG. 2, assummed with the dimmer voltage. The ballast 2 can also maintainbrightness if one of the lamps 4 burns out.

Referring back to FIG. 2, at start-up or turn-on, capacitor 144, whichis connected across pins 100 and 126, is discharged. When pin 100 comesup to its steady state or regulated voltage of 11.5 volts, capacitor 144pulls pin 126 to the regulated voltage. Capacitor 144 is then charged tothe regulated voltage of pin 100 by resistor 142 and the output ofamplifier 110. This causes the ballast to sweep in from the highestfrequency at start-up to a lower frequency as represented by the outputvoltage of amplifier 110. The lamps like this methodology because theyionize better at higher frequencies and the lower currents produced bytransformer 162. Essentially, this is a soft or gentle start for thelamps which preserves their fluorescent coatings and promotes longerlamp life.

Capacitor 144, once charged, also serves as part of a low frequencyfilter for the control system as connected between the output ofamplifier 110 and resistor 142 with resistor 142 to handle brightnessswitching transients. For example, when the dimmer control voltage ischanged by a user, the output of amplifier 110 changes almostinstantaneously. Similarly, if the lamp reference current developed incoil 132 changes, the output of amplifier 110 also changes almostinstantaneously. If the control system were to respond as quickly as itordinarily might to such changes, the lamps would flicker or flutteruntil the desired brightness was reached. To avoid this problem,capacitor 144, once charged, and resistor 142 form an RC circuit whichimposes a time delay on the signal applied to pin 126 and therebysmoothes the brightness transitions. Thus, the combination of capacitor144 and resistor 142, depending on where the control circuit is in theoperating cycle can act as a low pass filter when running or as adifferentiator at start-up.

The square wave voltage output of chip 104, as previously noted, isdependent on the voltage at pin 126. At initiation, pin 126 isrelatively high and thus the frequency of the voltage at pin 150 is alsohigh. Pin 150 goes to input pin 152 of chip 94. Internal to chip 94 aretwo buffers which place an output voltage on each pins 154 and 156 ofchip 94. These outputs are of the same frequency, but shifted 180degrees out of phase with each other. This has the effect of doublingthe voltage across the primary winding 162 a of pulse transformer 162.The internal buffers of chip 94 are driven by powerful drivers that arecapable of providing pulsed current flow in the order of 2 amps intocapacitive loads, the kind exhibited by a FET. This capability permits arelatively weak signal to be boosted so that the power FETs 158 and 160can be turned on very quickly. This has the effect of minimizingtransition losses, which are dependent on how fast the power FETs areturned on (in this application in approximately 40 nanoseconds). TheFETs are selected to have the lowest possible “on-resistance” orimpedance so that power losses through the FETs and in the ballast arekept to a minimum. Finally, chip 94 acts as a buffer between the lowpower CMOS implemented voltage control oscillator chip 104 and the powerFETs 158 and 160.

The outputs from driver pins 154 and 156 are a set of very closelymatched square waves whose edges are within 40 nanoseconds of each otherwith high pulsed drive (2 amp) capacity. The AC coupling effect ofcapacitor 164 permits the low impedance primary inductor 162 a to beeffectively connected to output pins 154 and 156. With this output, thecontrol circuit will drive the primary side of closely coupled pulsetransformer 162 a, the signal amplitude of which is effectively doubledat secondary transformer windings 162 b and 162 c to plus and minus 11volts by the out of phase output from pins 154 and 156. This effectivelyputs a 22 volt square wave across primary 162 a. On the secondary sideof transformer 162, this means that power FET 158 will have plus 11volts applied across its gate and source while power FET 160 will haveminus 11 volts applied across its gate and source. Since the FETs areselected with optimized values of minimal on-resistance when gate tosource voltage is greater than plus 5 volts and off-resistance ismaximal when gate to source voltage is less than minus 5 volts, they areeach turned on and off very quickly by the plus and minus 11 voltsapplied across their respective gate and source by the secondarywindings 162 b and 162 c respectively. This guarantees that the powerFETs 158 and 160 are turned on and off very quickly which minimizestransition losses.

The secondary windings 162 b and 162 c are out of phase with each otherby 180 degrees guarantying that the gate to source voltages generatedtherein that turn the power FETs on and off will also be out of phase bythat amount. However, the edges of the generated voltages are so sharpand fast that there is a possibility that the FETs could be on at thesame time, even if briefly permitting the 450 volts present at point 166to be conducted to ground. That would be unsafe and undoubtedly cause aproblem in the ballast or pose a threat to a user. Accordingly,inductors 168 and 170 are connected from one side of each secondarywinding to the source of the associated FET, as shown in FIG. 2, toimpose a slight delay and thereby establishing a safe zone and insuringthat the power FETs 158 and 160 are not on at the same time.

The center point 180 of the power FETs 158 and 160 is connected to theprimary side 178 a of a unique transformer 178 that will be describedhereinafter in greater detail. The on-off action of power FETs 158 and160 drives point 180 between 450 volts and ground. Capacitor 176provides AC coupling for primary winding 178 a. Capacitor 176, which isconnected to ground, charges to the middle of the voltage swing at point180 or to 225 volts. This effectively causes an AC voltage to beimpressed on primary winding 178 a that varies between 0 to 225 to 450volts. The diodes 172 and 174 are very fast and respectively serve toprotect the FETs 158 and 160 from any inductive kick that results fromabrupt voltage changes in the primary winding 178 a caused by the powerFETs shutting off.

Transformer 178 is an over-wound, current limiting type. When primary178 a turns on, transformer action causes voltage to be induced insecondary windings 178 b, 178 c, 178 d and 178 e. The voltage developedacross secondary winding 178 d during normal, steady state operation isapproximately 280 volts rms. The main power for lamps comes fromsecondary winding 178 d. Secondary windings 178 b, 178 c and 178 eprovide voltage to the lamp filaments 182, 184, 186 and 188. Thesecondary filament voltage developed by windings 178 b, 178 c and 178 eis 5 volts rms. As shown in FIG. 2, secondary winding 178 b is connectedto filaments 184 and 186, secondary winding 178 c is connected tofilament 182 and secondary winding 178 e is connected to filament 188.

At start-up, because it's over-wound, secondary winding 178 d goes toapproximately 470 volts rms, a voltage level that is needed to ignitethe lamps and cause ionization of their internal gas. At the same time,secondary windings 178 b, 178 c and 178 e provide approximately 9 voltsto the filaments. As previously noted for start-up, the drive frequencyis at its maximum value. At start-up, each of the capacitors 190, 192and 194 can be considered as shorted and the result is that the voltageacross the lamps from secondary winding 178 d is at a maximum to helpionize the gas within the lamps and cause gentle lamp ignition.Operating frequency is then at its highest possible value for thecontrol circuit portion and secondary 178 d current is at its lowestvalue, holding the filaments are at an elevated voltage level whichwarms the lamps and helps get them started by promoting electron flowfrom the filaments.

Since gas discharge lamps are easier to ionize at higher frequencies,the start voltage profile presented to the lamps promotes what is calleda “soft start.” The starting voltage for the lamps is predetermined tobe at an initial frequency of 100 KHz which is swept down to theoperating frequency of a non-dimming ballast or to the frequency setpoint corresponding to the feedback provided by the dimming switch (notshown) and associated dimming circuitry. At the higher operatingfrequency, less current is drawn in the secondary and that means thatless power is delivered to the lamps as a result of transformer 178action. This “soft start” results in significantly reduced flickeringand noise from the lamps during their start phase. In addition, thelower starting current reduces depletion of the phosphor on thesidewalls of the lamps thereby prolonging their life.

When the lamps start to draw current, secondary coil 178 d goes toapproximately 280 volts, a selected value that's typical for T8 type gasdischarge lamps (this value would be different for other types of gasdischarge lamps), due to the current limiting nature of transformer 178.The other secondary winding voltages and the filaments they areconnected to simultaneously drop to 5 volts for the same reason. Thefrequency starts decreasing to wherever the control point has been setat terminals 138 and 140.

Nominally, after start, secondary windings 178 b, 178 c and 178 e are at5 volts and stay at the level even as the frequency for control purposesdrops. However, the filament voltages are now dependent on the impedancepresented by capacitors 190, 192 and 194 to the respective filamentsthey are coupled to. The value of capacitors 190, 192 and 194 is toselect capacitive values that will drop actual filament voltages toabout 2.5 volts for full light or minimum control frequency or 5 voltsat 10 percent light, which corresponds to almost the maximum controlfrequency since at low light levels it is important to apply fullvoltage to the respective filaments to keep the lamps internally heatedand thereby avoid lamp flicker.

An equivalent, non-dimming electronic ballast 200 is shown in FIG. 4.The power supply portion of the non-dimming version of ballast 200, isidentical to the power supply portion of the dimming version of ballast2 shown in FIG. 2. The control circuit portion, of the non-dimmingversion of ballast 200 functions the same as the control circuit portionof the dimming version in all respects, except as follows. Differentialamplifiers 10b6 and 110 are removed from the control circuit of ballast2 together with the set of matched resistors 128 and terminal 138 and140, compare FIGS. 2 and 4. The end of resistor 142 that was connectedto the output of differential amplifier 110 is removed therefrom andconnected to input pin 127 of IC 104, compare FIGS. 2 and 4 once more.Pin 126 remains coupled between resistor 142 and capacitor 144. Atstart, capacitor 144 is discharged, effectively a short, which pulls pin126 to the top of its voltage range, insuring a maximum frequency outputvoltage on pin 150 to obtain the sweep in profile previously explainedthat the lamps favor. As capacitor 144 charges, the voltage to pin 126eventually diminishes to its minimum value, and the frequency of thevoltage at pin 150 drops linearly by the same percentage to the steadyoperating frequency. As preciously noted, the selected values ofresistors 122 and 124 determine the maximum and minimum frequencies forthe square wave voltage output on pin 150.

Also, in the non-dimming arrangement, the lamp current reference sensingcircuit, comprising sensing coil 132, resistor 134 and diode 136, arealso removed from the ballast 2 control circuit along with the lampfilament capacitors 190, 192 and 194, compare FIGS. 2 and 4. Theresultant non-dimming ballast 200 formed by removal of the aboveenumerated circuit elements and reconnection of resistor 142 directly toIC 104 is otherwise identical to dimming ballast 2.

The parameters of transformer 178 are selected to accommodate severalperformance factors including the power to be delivered to efficientlydrive lamps 4, the open circuit voltage required to initially turn onlamps 4 and the lamp current crest factor (the ratio of peak lampcurrent to the rms lamp current) which should be kept below 1.7. Inaddition, because of its current limiting capabilities, a short circuitor high current demand situation on the secondary side of transformer178 drops apparent power delivered by transformer 178 to its secondarywinding by an approximate factor of 10. There is an equivalent reductionof input power to the ballast as well.

The use of transformer 178, a frequency controlled, current limitingdevice, and frequency control of the ballast are the keys in providingan improved ballast. In addition, short circuit isolation is provided bythe transformer which isolates the load from the ultimate source ofpower and limits short circuit current to a small fraction of what itwould otherwise be.

Control is obtained by varying the voltage to input pin 126 of thevoltage controlled oscillator chip 104 to produce an output drivevoltage of essentially constant amplitude and variable frequency orholding the voltage (for a non-dimming version of the ballast) at aconstant value that produces a voltage at a predetermined constantfrequency. The net effect is that the current induced in the secondaryside of transformer 178 is directly dependent on the frequency of theapplied square wave. The use of such an arrangement limits current andvoltage as a function of frequency and negates any need to employ pulsewidth modulation and its associated resonate circuit to clean up thevoltage ripple. The present electronic ballast obviates that need whileproviding smoother, more efficient operating conditions.

In a loosely coupled transformer, the concept of leakage inductance is amathematical way to account for the less than theoretically possiblemagnetic coupling between its primary and secondary windings. FIGS. 5(a)and 5(b) depict loosely coupled transformers. The transformer coreconsists of two E-shaped, facing ferromagnetic members 502 and 504. Thecenter leg is defined by two openings in the core, shown in FIG. 5(b) as520 and 522. The center leg of the core may be constructed to include agap, as shown in FIG. 5(a), or the center leg may be gapless, as shownin FIG. 5(b). The primary and secondary coils (506 and 508,respectively) may be located at various positions on the perimeter ofthe core (see also 506 a and 508 a), as shown in FIG. 5(a).Alternatively, the primary and secondary coils 506 and 508 may belocated in a spaced apart relation along the center leg or member, andwound through two openings in the core as shown in FIG. 5(b).

In the embodiment of FIG. 5(a), the center portions of the E-shaped coreforms a shunt 510 which has an air gap 512 of predetermined, suitablewidth. Those having skill in this art will recognize that the gap 512 inshunt 510 can also be formed of different material than air; forexample, a dielectric material or even a fluid mixture of predeterminedcharacteristics. Further, it has been found through experimentation thatan appropriate leakage inductance can be achieved without a gap. Anexample of a gapless design includes a primary and secondary windinglocated on a center leg of a core, and spaced apart from each other.

A plastic bobbin 524 is preferably used to hold the windings. Wire iswound directly on the bobbin 524, which is then inserted on the centerleg. After insertion, the ferromagnetic core is bonded together, thuscapturing the bobbin 524. In the preferred embodiment, the bobbin has afirst section 526 for the primary winding 506 and a second section 528for the secondary winding 508. By separating the winding into differentsections, it has been found that the transformer will have relativelyhigh efficiency, while still generating an appropriate leakageinductance A separator 530 on the bobbin 524, preferably made of plasticand located between the first section 526 and the second section 528, isused to separate the primary winding 506 from the secondary winding 508.

In its usual operating state, case 2 as described hereinafter, theprimary flux path for transformer 500 is around the periphery asindicated by dotted line 514. Under short circuit conditions, primarymagnetic flux flow is through the shunt 510 and across the air gap 512,as shown by dotted line 516. Gap 512 can be varied as may be suitablefor various applications to shift the reluctance and response oftransformer 500 in accordance with the type of load and the controlschema being used therefor. It should be noted that the specific shapeof the transformer core is not limited to the depicted E-shape and thatan elliptical, circular or rectangular or other configuration can beemployed in a controller with the shunt placed in the core's outerperiphery or internal thereto as may be desirable. For those having aninterest in further and more specific details of transformers, includingthose of loosely couple transformers, reference should be made to theaforementioned Lee reference and in Transformers For Electronic Circuitsby N. R. Grossner, second edition, published 1983 by McGraw-Hill, NewYork, N.Y.

An equivalent model representation of the loosely coupled transformer500 is shown in FIG. 6. The inductance L_(p) of the primary side oftransformer 600 is considered to be entirely included in winding 602 andits resistive component R_(p) is considered to be entirely included inresistor 604. Thus, for purposes of modeling and analysis, the primarycoil is divided into a purely inductive component and a purely resistivecomponent. On the secondary side, winding 606 is considered to representthe secondary inductance L_(s). The load 608 is assumed to be purelyresistive for modeling purposes, consisting of the secondary windingresistance R_(s) and the load resistance R_(L)

The operating characteristics of this model and the coupling oftransformer 500 are determined as follows. Assuming a turns ratio of 1to 2, a step-up transformer, the inductance L_(p) of the primary and theinductance L_(s) of the secondary are measured when the secondary isopen circuited and short circuited. With the secondary of a test(loosely coupled transformer open, the apparent inductance of theprimary L_(po), was measured at 40 millihenries. With the secondary ofthe same test transformer shorted, the primary L_(ps) was measured at 40millihenries. The inductive ratios for this primary as given by:

Inductive Coupling Ratio=L_(po/)L_(·ps)  (2)

indicated a coupling ratio of 10. It should be noted that the same ratiowill be obtained by comparing secondary inductance under the samesecondary open and short circuited conditions, taking the transformer'sturn ratio into account. As a general rule of thumb, an inductivecoupling ratio below 10 is considered to be “loose,” while an inductivecoupling ratio above 30 is considered to be “tight.” As implemented, theinductive coupling ratio of a transformer should vary between 10 and 30.Alternatively stated, from 80% to 99% of the magnetic flux developed inthe transformer's primary winding, depending on the operatingcharacteristics of the transformer with respect to load requirements,should be flow through the transformer's secondary winding. This rangeis intended to encompass transformers with the electricalcharacteristics generally shown in FIG. 7.

Many variations of a leakage transformer are possible. Skilled artisanswill appreciate that several factors can affect the inherent leakageinductance of a transformer, including the shape of the core, theplacement of the windings, the size and shape of the gap (if any), andthe nature of the windings.

There are three basic cases of operation that can be used to define ordescribe the operating characteristics of transformer 500. For purposesof this description, it is again assumed that transformer 500 isoverwound so that a 2 to 1 turns ratio exists; that is, the transformeracts as a step-up device in which the input voltage V is doubled acrossthe secondary winding 606. For purposes of the simplified transformermodel to be discussed, it is further assumed that transformer 500 is“perfect” (no operating losses. R_(p)=0) in operation.

In the first modeled case, load 608 is considered to be very large,essentially an open circuit. In that instance, the secondary currentthrough load 608 and secondary winding 606 would be extremely small.Likewise, because transformer 500 is assumed to be “perfect,” theprimary current is also quite small. Thus, under open load conditions,transformer 500 acts to transfer a low amount of power from primary tosecondary in accordance with its windings turn ratio.

Skipping to case 3, load 608 is now considered to be extremely small,essentially a short circuit. In such a situation, the power delivered tothe load 608 is essentially zero (a very low or zero voltage multipliedby the secondary current) and I_(p), the current in the primary, isdetermined by the voltage applied to the primary winding and itsfrequency. Since transformer 500 is considered to be perfect, thecurrent in the primary is one half that of the secondary (with theassumed 2:1 turns ratio), and is, therefore, very low. Primary currentflow due to winding 602 is a function of the input voltage and frequencyand remains low as well as 90° out of phase with the voltage V_(p).Thus, as the secondary approaches a short circuit condition, apparentprimary current becomes low and drops toward zero amps.

In the intermediate case 2, where the secondary load 606 varies betweenopen circuit and short circuit conditions, the primary current I_(p) isdetermined by the voltage V_(p) impressed across the primary, itsfrequency and the short circuit inductance L_(ps) of the primary winding602. As shown in FIG. 7, at low frequencies, the winding current isrelatively high and is relatively low at high frequencies since, at anygiven frequency, the inductance of the transformer windings, andtherefore their impedance, varies inversely as a function of thefrequency of the impressed primary voltage.

For example, with load 606 at 1200 ohms, an input voltage of 225 voltsrms at a frequency of 100 kilohertz and an inductance L_(p) of 4millihenries and a primary resistance of 600 ohms, the primary currentI_(p) can be determined to be the primary voltage divided by itsimpedance Z_(p):

I_(p)=V_(p)/Z_(p)(where Z_(p)=2πF L_(p)−R_(p))  (3)

=225/(100×10³)(6.28)(4×10³)+(R_(p))=225/(2512+600)

=0.0723 amps or approximately 72 milliamps.

If the frequency in the intermediate case is increased to 200 kilohertz,primary current will drop to approximately 40 milliamps. If thefrequency is decreased to 50 kilohertz, primary current increases toapproximately 121 milliamps. Thus, changing the frequency of thetransformer's driving voltage permits control of power delivered to theload and as a limit for operating current, both as a function of thefrequency of the driving voltage applied to the transformer's primarywinding. The change in driving voltage effectively alters the operatingcharacteristics of the transformer by changing its inductive reactanceand impedance.

The resultant relationship between transformer current and frequency isshown in FIG. 7 in which a plot of primary current I_(p) and secondarycurrent I_(s) versus frequency of the impressed primary voltage isdepicted. It should be noted that the position of I_(s) with respect tothat of I_(p) will be a function of the turns ratio of the transformer'sprimary and secondary windings.

The ability to control the characteristics of a loosely coupledtransformer where its winding inductance, and therefore its impedance,can be varied as a function of frequency can be utilized in a controlleror regulator as previously described in connection with ballast 2. Asimplified general schematic diagram of such a controller is shown inFIG. 8. Line voltage V at a level and frequency commonly available isprovided to a rectifier circuit 802 which rectifies that input as may beneeded by the other circuitry and load. The rectified and possiblyboosted or reduced voltage, to the extent such boosted voltage may beneeded in a specific application, is supplied to a voltage conditioningand control circuit 804. Control circuit 804 forwards a referencevoltage signal to the frequency control circuit 806, which adjusts thefrequency of its output in accordance with that reference, subject toany chancres in that output frequency, as may be called for and requiredby the load 810 though its feedback loop 812.

The output from the frequency control circuit block 806 is fed tocontrol circuit 804 where it is used to produce a desired voltage type(e.g—square wave, varying pulse train, haversines, etc.) that isforwarded to transformer driver circuit 808. The transformer driversthereby created in block 808 are used to generate drivers fortransformer 500 which is coupled to the load 810 whereby power isdelivered thereto. It will be readily apparent, as explained above, thatcontrol of the transformer's operating characteristics can be suitablymade dependent on the frequency of the voltage impressed upontransformer 500's primary winding. Furthermore, it can readily be seenfrom the relationship depicted in FIG. 7 and the controller schematic ofFIG. 8, and as demonstrated in connection with the dimming version ofballast 2, that load feedback can be used to appropriately vary thefrequency of the voltage impressed on the transformer to vary itswinding currents and thereby achieve regulated and/or controlledtransfer of waveforms, power or load control.

An example of how a controller 900 can be used is illustratedschematically in FIG. 9. In the FIG. 9 arrangement, a thermoelectriccooler/heater 902 is controlled by the application of appropriatevoltage from loosely coupled transformer 500 and rectifier circuit 904.As external temperature conditions change or as a user desires, feedbackto or a set point signal from temperature controller 906 is used toadjust the output of frequency controller 908 to thereby change thefrequency of the driver voltage impressed upon transformer 500 in amanner previously discussed in connection with the operation ofelectronic ballast 2.

Another example of how a controller 1000 can be employed isschematically shown in FIG. 10. In the FIG. 10 arrangement, pulse drivercircuitry 1002 is adapted to deliver a pulse train of varying frequencyto a remote receiving circuit or load 1004 through loosely coupledtransformer 500. Voltage applied to transformer 500 is set so to thelowest or fundamental frequency of pulses delivered by the pulse drivercircuitry 1002 so that the maximum current drawn would be limitedthereby. In the event of overload or malfunction in the secondary,feedback to shift the operating frequency point upwardly can beadvantageously employed co limit current.

Yet another example of how a controller 1100 can be utilized isillustrated in FIG. 11 wherein source voltage from a public powerprovider 1102 is passed to a substation or industrial complex load 1104via loosely coupled transformer 500. Alternatively, the load 1104 couldconsist of a motor or set of motors which are periodically placed on ortaken off line. Transformer 500 could be a step up, step down or unitywound transformer, but it can be adapted by frequency control of itscharacteristics so that heavy output load or a short circuit could notpull down or reduce the source voltage to other users.

Dimming Ballast For Parallel Gas Discharge Lamps:

Referring now to FIG. 14, another alternative embodiment of the dimmingballast in accordance with the present invention is shown. This uniqueembodiment utilizes four loosely coupled transformers T2, T3, T4, and T5each having its primary winding connected in parallel to the output ofpower FET 158 and power FET 160. The secondary outputs of looselycoupled transformers T2, T3, T4 and T5 go to connector J3. Connector J3connects to a plug P3 which in turn connects to four loads such asfluorescent lamps 1501, 1502, 1503, and 1504 shown in FIG. 15.

The power FETs are driven through transformer T1, i.e., transformer 162of the dimming electronic ballast in FIG. 2. Accordingly, the power andload circuits connected to the secondaries of transformer 162 in FIG. 2are replaced by the power and load circuits connected to the secondariesof transformer T1 in FIG. 14 to make up the embodiment of the presentinvention shown in FIG. 14. A further change between the embodiment inFIG. 14 and the embodiment in FIG. 2 is that no delay time inductors 168and 170 in FIG. 2 are connected to the sources of FETs 158 and 160.These inductors were used to provide a dead time between the switchingof the FETs to ensure that they both were never on simultaneously. Inthe alternative embodiment in FIG. 14, a small delay or phase shift (notshown) would be inserted between the two inputs 152 (FIG. 2) into thebuffers in integrated circuit 94. As a result the square wave outputfrom the secondaries of transformer T1 or 162 would have a slight deadtime at each zero voltage crossover in the square wave. This ensuresthat FETs 158 and 160 are never on at the same time. The diodes 172 and174 across the FETs are very fast and respectively serve to protect FETs158 and 160 from any inductive kick that results from abrupt voltagechanges in the primary windings caused by the power FETs shutting off.

Terminals 3 of transformers T3 and T5 are connected to the source of FET158. Terminals 3 of transformers T2 and T4 and terminals 6 oftransformers T3 and T5 are connected to the source of FET 160 throughcapacitor 176. The center point 180 of the power FETs 158 and 160 isconnected to terminals 6 of transformers T2 and T4 and terminals 3 oftransformers T3 and T5. This arrangement results in transformers T4 andT5 always being 180 degrees out of phase with each other, and thereforethe voltage produced between them is always a net zero. Similarly,transformers T2 and T3 are 180 degrees out of phase, and therefore thevoltage produced between them is also zero.

The on-off action of power FETs 158 and 160 drives center point 180between 450 volts and ground. Capacitor 176 provides AC coupling forprimary windings of transformers T2, T3, T4, and T5. Capacitor 176,which is connected to ground, charges to the middle of the voltage swingat point 180 or to 225 volts. This effectively causes an AC voltage tobe impressed on the primary windings that varies 225 volts above andbelow the voltage swing point of 225 volts. In addition, the maximumvoltage between any two output leads from the secondary windings willnot exceed 225 volts rms.

The embodiment in FIGS. 14 and 15 is a true parallel configuration forgas discharge or fluorescent lamps. Therefore, when one lamp of the fourfails, the remaining three continue to run normally, regardless ofwhether the failed lamp is shorted or opened.

In FIGS. 14 and 15, each of the transformers T2 through T5 is a looselycoupled transformer as described above, in which the primary tosecondary turns ratio is 1:1. The primary has 363 turns and the powersecondary (terminals 2 and 5) has 363 turns. Typically, the filamentsecondary windings of each transformer T2, T3, T4, and T5 each has 16turns and include a 0.1 microfarad capacitor in series. The value of thecapacitor is chosen so that at high frequencies during startup thecapacitor is a low impedance so that current flows through the filamentof the lamp. At lower frequencies during lamp operation the impedance ofthe capacitor is high to limit filament current to effectively place thefilament windings out of the circuit when the lamps are running normallywith the full running voltage provided by the main windings (terminals 2and 5) of each transformer. For example, during startup, lamp 1501receives filament current from terminals 1 and 2 of transformer T5through capacitor C17 and from terminals 4 and 5 via capacitor C21. Themain secondary winding between terminals 2 and 5 provides the maincurrent through the lamp 1501 when the gas in the lamp 1501 is ionizedand a plasma formed at full power. The filament current may not beneeded during full power operation and capacitors C17 and C21effectively limit the current to the filament windings in thiscondition. The full secondary current is provided by the winding betweensecondary terminals 2 and 5 of each transformer.

In addition, secondary terminal 5 of transformer T5 is connected tosecondary terminal 2 of transformer T4, secondary terminal 2 oftransformer T3 is connected to secondary terminal 5 of transformer T4,and secondary terminal 2 of transformer T2 is connected to secondaryterminal 5 of transformer T3. The 363 turn secondary winding betweenterminal 2 and 5 of transformer T5 is connected across the gas dischargevolume of lamp 1501 through pins 2 and 3 of the jack J3 and plug P3. The363 turn secondary winding between terminal 2 and 5 of transformer T4 isconnected across the gas discharge volume of lamp 1502 through pins 3and 5 of the jack J3 and plug P3. The 363 turn secondary winding betweenterminals 2 and 5 of transformer T3 is connected across the gasdischarge volume of lamp 1503 through pins 5 and 7 of the jack J3 andplug P3. Similarly, the 363 turn secondary winding between terminals 2and 5 of transformer T2 is connected across the gas discharge volume oflamp 1504 through pins 7 and 9 of jack J3 and plug P3.

Transformers T2, T3, T4,and T5 are all wound with the correspondingprimary and secondary winding ends correspondingly connected as is shownby the dots adjacent primary terminals 3 and secondary terminals 1, 2,and 5 in each of the transformers T2, T3, T4, and T5 in FIG. 14. Eachlamp 1501, 1502, 1503, and 1504 is driven by its own transformer. Inaddition, with the connection arrangement shown in FIGS. 14 and 15, onlyone lamp voltage is applied to each lamp so that failure or an opencircuit in one lamp circuit will not affect the operation of any otherlamp.

While the preferred embodiment of the invention in FIGS. 15 and 16 showsfour parallel gas discharge lamps, any number of lamps could be placedin parallel. It is advantageous to add lamps in pairs so that oppositephase transformers may balance out the voltage between them. The numberof lamps is only limited by the power available from the power supplyand through the power FETs.

In another embodiment of the invention, shown in FIG. 16, a couplingtransformer T6 is placed between the output center power point 180 andthe AC coupling capacitor 176. The coupling transformer T6 couples thepower output to a coaxial cable CA. The cable CA carries the highfrequency power signal to the primary windings of the loosely coupledtransformers T2, T3, T4 and T5. The ballast control circuitry of FIG. 14could be located in one cabinet and the loosely coupled transformers T2,T3, T4 and T5 and the lamps 1501 through 1504 shown in FIG. 15 simplylocated at the light fixture itself. The coaxial cable CA facilitateslocating the ballast control 60 feet or more away from the actuallocation of the lamps. This arrangement would be very convenient from amaintenance and control perspective.

Further, the phrase “overwound loosely coupled transformer” as used inthis description does not refer to the turns ratio. An overwound looselycoupled transformer simply means that the open circuit voltage at thesecondary winding is higher than the operating voltage of the devicebeing driven by the loosely coupled transformer. The actual operatingvoltage of the device being driven at the secondary of the transformeris determined by the leakage inductance of the transformer. At theoperating frequency the overwound loosely coupled transformer is acurrent source.

It will be apparent to those of skill in the appertaining arts thatvarious modifications can be made within the scope of the aboveinvention. For example, four lamps are illustrated in the embodimentshown in FIGS. 14 and 15. Additional loosely coupled transformers couldbe added along with additional lamps connected in a similar fashion toproduce parallel arrangements of five or more lamps. Similarly, a fewernumber of lamps may be utilized, e.g. three, in which case, one of thefour transformers could be eliminated. Accordingly, this invention isnot to be considered limited to the specific examples chosen forpurposes of disclosure, but rather to cover all changes andmodifications which do not constitute departures from the permissiblescope of the present invention. Having thus described my invention, whatis desired to be secured and covered by Letters Patent is presented inthe appended claims.

What is claimed is:
 1. A controller for regulating power delivered to aplurality of gas discharge lamps, said controller comprising: a powersupply circuit; a voltage controlled oscillator power circuit connectedto the power supply circuit, wherein said voltage controlled oscillatorpower circuit has a input and an adjustable frequency output; and aplurality of loosely coupled transformers each having a primary windingand at least one secondary winding, wherein the primary windings areoppositely electrically connected to the adjustable frequency output,and wherein the at least one secondary winding is electrically connectedto supply current to a gas discharge lamp, wherein said current isresponsive to said adjustable frequency output.
 2. The controlleraccording to claim 1 wherein said adjustable frequency output isconnected to an output stage connected to each of the loosely coupledtransformers.
 3. The controller according to claim 2 wherein two of saidtransformers have primary windings connected in a first direction tosaid output stage and the other two of said transformers have primarywindings connected to said output stage in a direction opposite to saidfirst direction.
 4. The controller according to claim 3 wherein eachtransformer has a pair of filament secondary windings and a main windingtherebetween wound in a direction corresponding to said primary windingof each said transformer.
 5. An electronic ballast for supplying powerto a plurality of gas discharge lamps each having a brightness, saidelectronic ballast comprising: a power supply circuit; a voltagecontrolled oscillator connected to the power supply circuit, whereinsaid voltage controlled oscillator has an input and an adjustablefrequency output; and a plurality of loosely coupled transformers eachhaving a primary winding and a plurality of secondary windings, whereinthe primary winding of each is electrically connected in parallel to theother loosely coupled transformers and to the adjustable frequencyoutput, and wherein each of the secondary windings is electricallyconnected to one of the lamps and one secondary winding of eachtransformer is oppositely connected in parallel to another secondarywinding of another of said transformers, wherein said brightness isresponsive to said adjustable frequency output.
 6. A method forregulating electrical current utilizing a loosely coupled transformerconnected to a plurality of parallel connected gas discharge lamps, saidmethod comprising the steps of: supplying power to a voltage controlledoscillator to thereby create an output having a frequency; supplying theoutput to the loosely coupled transformer to thereby supply current to aload; and adjusting the frequency of the output to thereby adjust thecurrent delivered to each of the lamps according to reactance in theloosely coupled transformer.
 7. An electronic ballast for supplyingpower in parallel to a plurality of gas discharge lamps each providingillumination, said electronic ballast dimming the illumination from eachlamp and comprising: a power supply circuit; a controlled oscillatorpower circuit connected to the power supply circuit, wherein said powercircuit has an input and an adjustable frequency output; a plurality ofloosely coupled transformers connected in parallel to said adjustablefrequency output, each of said transformers having a primary winding anda secondary winding wound in a first direction, wherein one of saidprimary windings is connected to said adjustable frequency output insaid first direction and a primary winding of another of saidtransformers is connected to said adjustable frequency output in adirection opposite to said first direction whereby voltage developed insaid secondary winding of said one of said transformers has a polarityopposite to voltage developed in said another of said secondary windingsof said other transformer; and each of said secondary windings connectedto one of said gas discharge lamps whereby the illumination provided byeach gas discharge lamp varies with the adjustable frequency output ofsaid power circuit.
 8. The ballast according to claim 7 wherein saidtransformers each have at least two secondary windings and eachtransformer has at least one secondary winding connected to anoppositely wound secondary winding of another of said transformers.